The present invention relates in general to RF oscillator/detectors of the type that are used for conducting electrical measurements of particles (e.g., blood cells) contained in a carrier fluid in a flow cytometer system. The invention is particularly directed to a new and improved solid state RF oscillator-detector circuit, that employs a dual junction field effect transistor (JFET)-based Hartley RF oscillator, having a relatively low Q tank circuit, that is coupled to the flow cell by an impedance-matching transformer.
As an adjunct to the diagnosis and treatment of disease, the medical industry commonly employs various types of particle flow cytometers, such as that diagrammatically illustrated at 10 in FIG. 1, to analyze particles in a patient""s body fluid (e.g., blood cells). For analyzing a patient""s blood, for example, a whole blood sample is initially diluted with a saline solution, lysed to explode all the red cells, and then stabilized to return the remaining white cells to their original size.
The prepared blood sample is then placed in a sample holding chamber 12, and a stream of the blood sample is conveyed along a flow channel 11 from the holding chamber 12 through a restricted orifice or aperture 14, that allows particles to be counted one at the time, and into a receiving chamber 16. Via electrodes 21 and 23 that are respectively coupled to either end of the flow cell""s holding chambers (holding chamber 12 and receiving chamber 16) a DC electrical field for measuring the displaced volume of each particle and an RF field for measuring the density of each particle passing through the aperture 14 are applied to the flow cell 10 by way of an oscillator-detector circuit 17, which is preferably configured as a Hartley oscillator (although other oscillator architectures may also be used).
As particles pass through the flow cell orifice 14, they introduce changes in the resistance of the orifice in proportion to their size or volume. These changes in resistance are reflected as DC voltage pulses at the electrodes 21 and 23. The density or opacity of the blood cells is associated with changes in reactance of the flow cell aperture 14. By coupling the electrodes 21 and 23 of the flow cell 10 in parallel with the resonance (LC tank) circuit of the RF oscillator-detector circuit 17, changes in the reactance of the flow cell are reflected as a corresponding change in the operation of the RF oscillator, which is measured by means of an RF pulse detector/demodulator.
For non-limiting examples of U.S. Patent literature detailing conventional electronic tube based flow cell RF oscillator detector circuits, attention may be directed to the U.S. Patents to Coulter et al, Pat. No. 3,502,974: Groves et al, Pat. No. 4,298,836; Groves et al, Pat. No. 4,525,666; and Coulter et al, Pat. No. 4,791,355.
Now although a tube-based flow cell measurement circuit of the type shown in FIG. 1 is effective to provide an indication of both particle size and density, it suffers from a number of problems which are both costly and time-consuming to remedy. A fundamental shortcoming is the fact that it was originally designed as and continues to be configured using relatively old electronic tube components. This potentially impacts component availability, as the number of manufacturers of vacuum (as well as gas filled) electronic tubes continues to decline. In addition, the effective lifetime of a newly purchased and installed tube in the RF (Hartley) oscillator is not only unpredictable, but experience has shown that the effective functionality of most tubes within the Hartley oscillatorxe2x80x94detector circuit is very limited, (even though a tube tester transconductance measurement shows a tube to be good). At best a tube can expect to last somewhere in a range of three to nine monthsxe2x80x94and typically involves on the order of two repair/maintenance service calls per year per flow cell.
While it might seem that a straightforward solution to the tube aging problem would simply involve replacing the electronic tube (e.g., triode) with a solid state device, such as a bipolar transistor, MOSFET, JFET and the like, such is not the case. Investigation by the present inventors has revealed that, in order to exhibit the sensitivity necessary to successfully function as a detector, the tube must operate over a relatively narrow, steep sloped region of its plate current versus plate voltage relationship, shown at 27 in the triode characteristic of FIG. 2.
It has been found that the relatively short mean time before failure (MTBF) of a conventional electronic tube-based flow cell measurement circuit is due to the fact that, as the tube ages, the slope of its plate current versus plate voltage characteristic at VGRID=0 falls off quickly, and thereby degrades the tube""s sensitivity to the extent that it no longer effectively functions as a detector, even though it may continue to operate as an RF oscillator.
If one considers the active device""s (tube or JFET) operating range sensitivity (plate or drain voltage vs. grid or gate voltage) as a measure of transconductance (gm) dependence, from a comparison of the respective characteristic curve sets shown in FIGS. 5A (triode) and 5B (JFETs), it can be readily seen that a JFET provides a considerable improvement over a tube.
Typically, for a triode, this becomes 300v/0.1v=3000:1 vs. for a JFET 20v/0.1v=200:1. This is very important, given the small change in grid/gate voltage for a disturbance caused by the blood cell in the flow cell. Thus, an electronic tube will see a times fifteen degradation over a JFET for the same grid/gate voltage change, which makes the tube very dependent upon it""s transconductance gm. A small decay in the tube""s gm will then result in complete loss of detection capability. Thus, simply reconfiguring a conventional tube-based Hartley oscillator out of solid state components will not necessarily solve the problem.
In accordance with the present invention, the discovery of the above-discussed sensitivity-dependent slope limitation requirement has led the present inventors to design a new and improved solid state-based Hartley oscillator-configured flow cell detection circuit, that not only solves the tube-aging problem, but provides substantially improved performance. As will be described, the oscillator-detection circuit of the invention employs a pair of JFETs as its principal active devices (respectively operating in Class C and Class AB mode), which enables the circuit to achieve near zero noise operation with a very high VDS vs. IDS slope at a VGS=0 volts.
Advantageously, JFETs are inherently noiseless, except for the thermal noise intrinsic with channel resistance between the drain and the source. In the operation of the oscillator/detector, it is very easy to be misled as to the value of rms noise level seen at the detector output. The circuit noise that is coupled to the detector output is primarily related to the conduction time of JFET channel resistance. The shorter conduction time, reduction of channel resistance, or reduction of channel current, the lower the effective noise.
As will be described, operation with two JFETs in different class modes helps reduce the noise floor. A low current in the Class AB JFET stage in combination with low channel resistance allow for a lower noise floor. When the Class C JFET stage switches on, then only for that time is the additional channel device a noise source. The tradeoff is conduction time vs. the product of conduction current and conduction resistance.
In accordance with a preferred embodiment of the invention, a pair of parallel-coupled JFETs having different transfer functions, in particular different pinchoff VGS and max IDSS characteristics, are employed as the principal active element of the RF oscillator. As pointed out briefly above, there are two modes of operation that occur in both a JFET and a triode tube, as shown in FIGS. 5A and 5B, respectively. As far as RF mode operation is concerned, both devices are operated in their linear saturated regions with the RF load lines.
However, for the detection process, both devices operate in their square-law regions, as shown in the pulse load line. This is not intuitively evident from a circuit simulation, as only the RF region is operative and the""simulation models do not include the square-law region. Operation in the saturation region cannot develop any detectable change due to a perturbation in the loading by a cell. The detection process operates near Vgs=0v and Vgc=0v, where the highest slope in the square-law region occurs. Both circuits are biased for the saturated region to support RF generation.
There has been considerable study on the temperature effects of the JFET to detection stability. A single JFET device can be biased such that it can be made substantially independent to effects of temperature. However, this biasing condition causes the JFET to be operated, such that Vgs is quite far away from Vgs=0v. The net result is that the oscillator will not function as a detector. While it is possible to cause the biasing to change as a result of temperaturexe2x80x94which stabilizes the JFETxe2x80x94the net result is that the correction activity introduces a noise source, that limits its usefulness.
With a pair of JFETs operating with different parameters, each device will be set at a different temperature, which leads to problems with temperature stability. While it may be possible to selected two JFET devices such that they will cancel out each other""s temperature curves, this is not a viable solution from a manufacturing perspective. As a result, it is preferred to install the two JFETs and an associated current mirror in a temperature control chamber. This provides the circuit designer with considerable latitude in the choice of JFETs, as only the detection process needs to be considered.
In a preferred embodiment of the invention, the respective parallel-connected source-drain paths of the two (Class C, Class AB) JFETs are coupled between a DC voltage supply node and a center tap of a primary winding of a flow cell impedance-matching, ferrite core toroid transformer. This transformer also forms an inductive component portion of a relatively low Q resonator circuit that sets the fundamental resonant RF frequency of the oscillator. The frequency of the low Q tank circuit can be adjusted by a variable capacitor.
The transformer""s primary winding is coupled to parallel connected gates of the JFETs through a gate input circuit, that includes a DC battery (resistor-capacitor) path for increasing gain as a bootstrap impedance feedback at low frequencies, and a parallel capacitor path that effectively bypasses the battery at RF frequencies.
The transformer allows the required gate biasing resistance to be matched to the load presented by the flowcell. By matching to the flow cell load is meant that the low Q tank circuit""s transformer is power-matching the RF oscillator to the flow cell for optimum detection sensitivity. This is not meant to imply that the impedance of the flow cell is being matched to that of the RF oscillator. In a tube-based circuit of the prior art, the grid bias resistance can be very high, for example on the order of one megohm, which allows two things to occur. First, the grid bias resistance has no loading impact. on the tank circuit. Secondly, the tank circuit can have a very high Q (e.g., on the order of 120).
Using a transformer to enable a relatively low gate resistance to bias the JFET requires two parameters from the tank circuit: the tank Q must below (e.g., between 8 and 20), as gate resistance dominates the loading, and a step-up secondary winding provides matching between the lower impedance of the JFETs and the higher impedance of the flow cell. In addition, the RF voltage applied across the flow cell can be considerably higher than could be tolerated by the JFETs directly.
More particularly, the RF voltage presented across the flow cell itself is approximately what is seen at the JFETs gates. However, an AC voltage divider is formed between the transformer secondary winding and the flow cell with a capacitor. The capacitor forms part of the impedance matching between the flow cell, a coaxial feed to the flow cell, and the RF oscillator. Since a DC current is also presented to the flow cell to measure the volumetric displacement of a particle, the secondary winding of the transformer is AC-coupled to the flow cell. The capacitor serves to match the Rf oscillator to the flow cell and its coaxial feed, while blocking the DC current of the volumetric measurement. Within reason, the higher the applied RF voltage across the flow cell, the more sensitive the RF oscillator/detector becomes to a dielectric impedance change caused by the presence of a particle (blood cell) in the detection aperture.
The primary winding of the low Q tank circuit""s transformer is further coupled to a current sink compliance voltage load sensing node of a current (sink) mirror circuit. The current mirror circuit is operative to cause the RF oscillator to function as a load detector, by multiplying current variations by a synthetic high resistance, and is configured to maintain a constant output impedance throughout changes in compliance voltage. To optimize its functionality, the slope of the collector current vs. base voltage characteristics of its two bipolar transistors is relatively shallow, so that with load changes the output impedance will remain effectively constant and high.
The current mirror is coupled to a bypass capacitor which provides both a low impedance path to ground for the RF signal, and serves as an energy storage device for ensuring a good transient response for the current mirror circuit. The bypass capacitor serves to capture a change in RF oscillator load due to a particle passing through the flow cell aperture. The value of the bypass capacitor is chosen to match the RF impedance seen looking into the tank transformer. Thus, the value of the capacitor will have the same RF impedance as that of the tank winding. This matching of the RF impedances will yield the maximum detected load change signal.
As pointed out above, the RF oscillator employs both a Class C JFET and a Class AB JFET. For optimum operation in Class C the conduction angle is 153 degrees. Class AB causes the conduction angle to be increased to a value between 200 and 300 degrees. Since there is no steady state conduction of either JFET, the JFETs may be considered to be operating as current pumps rather than as linear devices. Each JFET injects a current pulse simultaneously with the cyclic swing of the tank circuit. The Class AB JFET has a higher pinchoff voltage and lower max Idss than the Class C JFET. As a consequence, the Class AB JFET injects a smaller current pulse but of longer duration into the tank circuit than the Class C JFET.
The Class C JFET injects a power pulse that rapidly ramps up the gain of the loop much higher than the other JFET could achieve. Since noise is a function of current and time into an impedance, then if the power pulse is shorter than. the average, the amount of noise energy is reduced. What is effectively achieved is a tradeoff between that required to sustain operation as an RF oscillator and what is required to function as a load change detector. The change in pulse current is coupled to downstream amplification circuitry.
In operation, a DC current source delivers a prescribed current coupled by the flow cell interface circuit to a flow cell electrode, to produce a DC electrical field for measuring the size of each particle passing through the flow cell""s detection aperture. A disturbance in this DC electric field due to a particle is reflected by a change in compliance voltage of the current source. When particle size within the aperture increases, the aperture resistance will also increase, increasing the current source compliance voltage, as the RF oscillator requires less current pulse injection to maintain RF amplitude. To detect a change in particle opacity or density, the nominal RF frequency is coupled by the transformer secondary through the interface circuit to the flow cell. The presence of a particle in the flow cell aperture causes a change in flow cell reactance, as the resistance and capacitance of the aperture are effectively part of the resonant circuit.
Although the Q of the transformer-configured tank circuit will increase slightly due to the presence of a particle in the aperture, this does not have nearly the impact on the JFET oscillator""s operation as in a tube design. In a high Q tank circuit, the presence of a particle causes the oscillator""s frequency to shift upwards towards the Q peak of the tank. The closer the oscillator frequency approaches that of the Q peak, the less pulse injection current is required to maintain the oscillator""s voltage amplitude.
For the case of a low Q tank, there is little change in frequency due to the presence of the particle, as there is no significant tank resonance frequency. Still, there will be a reduction in loading and the JFET will need to inject less of a current pulse into the tank, to maintain the oscillator""s amplitude. As a consequence, a low Q tank design responds almost exclusively to the real resistance change caused by the loading of a particle. A high Q tank, however, is very sensitive to both the real and reactive load changes, as the reactive change causes significant changes in the .oscillator""s frequency. This is an important issue as the dual JFET detector of the invention responds only to the power loading changes caused by a particle, which results in better small particle linearity. This improvement in linearity is seen mostly in particles that are smaller than five microns in diameter.